The inherent offset of an integrated amplifier is on the order of millivolts (mV). To process baseband signals in the microvolt (μV) or less region, a means of reducing the offset of the amplifier is required. It is also desirable to reduce the level of the low frequency noise without using large input transistors. One method for correcting the inherent offset is shown in FIG. 1.
FIG. 1a illustrates a prior art feed-forward amplifier for correcting the inherent offset of the amplifier. The feed-forward amplifier includes a main differential amplifier 102, an auxiliary amplifier 103, a second stage amplifier 104 and its compensation capacitor 105, and an output amplifier 106. The main differential amplifier 102 has two sets of inputs. The first set of inputs is 110, and the second set of inputs is 112, which is coupled to a first hold capacitor 114. Similarly, the auxiliary amplifier 103 has two sets of inputs. The first set of inputs is 111, and the second set of inputs is 119, which is coupled to a second hold capacitor 120. Both the main amplifier 102 and the auxiliary amplifier 103 are controlled by the switches SW1 115, SW2 116, SW3 117, and SW4 118.
The feed-forward amplifier has two modes of operation. In the first mode, the auxiliary amplifier 103 is cascaded with the main amplifier 102. In the second mode, the main amplifier 102 is used alone. In the first mode, the forward gain of the composite amplifier is the product of the gain of the main amplifier 102 and the auxiliary amplifier 103. In the second mode the gain is just the gain of the main amplifier 102.
When the feed-forward amplifier is used in an application, the gain developed may be represented by
  Av  =      Av          1      +              Av        ⁢                                  ⁢        B            where Av is the forward gain of the amplifier and B is the feedback attenuation. This gain can be rearranged to
      1    B    ×      1          1      +              1                  Av          ⁢                                          ⁢          B                    where the first term represents the ideal gain for the circuit and the second term represents the error due to the finite loop gain.
In the feed-forward amplifier, the error has a time varying component as the amplifier cycles between the first mode and the second mode. Multiplying by a periodic time varying gain is a well-known mixing process used in many radios. In this case, the mixing operation produces sidebands to the desired signal offset by the auto-zero frequency. These sidebands can adversely impact the operation of the feed-forward amplifier when the input frequency is very close to the auto-zero frequency and the side band appears at or around the frequency of direct current (DC). This problem described above is commonly referred to as the inter-modulation problem.
FIG. 1b illustrates another prior art circuit for correcting the inherent offset of the amplifier. As shown in FIG. 1b, the circuit includes a first differential amplifier (amplifier A) 122, a second differential amplifier (amplifier B) 123, a second stage amplifier 124 and its compensation capacitor (Cc) 125, and an output amplifier 126. The first differential amplifier 122 has two sets of inputs. The first set of inputs is 130a and 130b. The second set of inputs is 132 and 133, which are coupled to a first reference voltage (Vref_a) and a first hold capacitor (CH1) 136 respectively. The input/output (I/O) of the first differential amplifier is controlled by the switches SW1 140, SW2 142, SW6 152, and SW7 154. Similarly, the second differential amplifier 123 has two sets of inputs. The first set of inputs is 130a and 130b. The second set of inputs is 134 and 135, which are coupled to a second reference voltage (Vref_b) and a second hold capacitor (CH2) 138 respectively. The I/O of the second differential amplifier is controlled by the switches SW3 146, SW4 148, SW5 150, and SW8 156.
The circuit has two modes of operation. In the first mode, input amplifier A 122 is being used in the signal path and input amplifier B 123 is being auto-zeroed. In this mode, the switches SW2 142, SW3 146, SW6 152, and SW8 156 are closed and switches SW1 140, SW4 148, SW5 150, and SW7 154 are open. In the second mode, input amplifier B 123 is being used in the signal path and input amplifier A 122 is being auto-zeroed. In this mode, SW1 140, SW4 148, SW5 150, and SW7 154 are closed and switches SW2 142, SW3 146, SW6 152, and SW8 156 are open. The first mode and second mode are operated alternatively.
When either input amplifier A or input amplifier B is being used in the signal path, the main inputs 130a and 130b are connected to the input pins of the circuit through the switches SW2 142 or SW4 148 respectively. The offset of an input amplifier is the sum of the inherent offset of the amplifier and the voltage difference applied to the auxiliary inputs (132 and 133 for amplifier A, and 134 and 135 for amplifier B) which were set during the prior phase. The output of the input amplifier is coupled through either SW5 150 or SW6 152 to the second stage amplifier 124 and its compensation capacitor Cc 125 for additional gain and frequency shaping. The output of the second stage amplifier is buffered by the output amplifier 126 to drive an output load.
When either input amplifier A 122 or input amplifier B 123 is in the auto-zero mode, the main inputs 130a and 130b are shorted by either SW1 140 or SW3 146 and the output is connected to the auxiliary inputs. The amplifier will reach a steady state where the voltage difference applied to the auxiliary inputs cancels the inherent offset of the input amplifier to within an accuracy determined by the finite gain of the loop. This voltage will be held by either the hold capacitor CH1 136 or CH2 138 when SW7 154 or SW8 156 is opened. The auto-zeroed input amplifier is then used in the signal path.
There are a number of issues with the circuit of FIG. 1b. First, errors may result from the charge injection of the sample switch SW7 154 onto the first hold capacitor CH1 136 and the charge injection of the sample switch SW8 156 onto the second hold capacitor CH2 138. The offset from the charge injection may be represented by
            1      Kg        ×                  Q        inj                    C        Hold              ,where Kg is the gain from the main inputs 130a and 130b divided by the gain from the auxiliary inputs (132 and 133 for amplifier A, and 134 and 135 for amplifier B), Qinj is the charge injected by the sample and hold switch (on the order of half the gate capacitance times the clock swing), and Chold is the hold capacitor (CH1 136 for amplifier A, and CH2 138 for amplifier B). In older low-cost processes, the charge injection may be as high as 20 femto-Coulomb (fC). In newer processes, the charge injection may drop to as low as 0.25 fC, but the voltage capabilities in such new processes tend not to be adequate for the high dynamic range applications. The size of the hold capacitor 114 is typically limited, which restricts its capacitance to no more than several 10 s of pico-Farad (pF) to minimize expensive chip area. The relative gains from the two sets of inputs are about the same, the Kg is about one. Thus for older processes with a Qinj of 20 fC, a hold capacitor of 20 pF, and a Kg of 1, the input offset may be about 1 mV, which represents no significant improvement over the uncorrected offset. For newer deep sub-micron processes, the offset may be 12.5 μV, which is insufficient for applications that require signals in the microvolt region.
In addition, there is an offset error resulting from the finite gain of the amplifier. When the amplifier is in auto-zero mode, the input differential is zero since the inputs 130a and 130b are shorted together. However the output of the amplifier is at a voltage that is determined by the auxiliary inputs, not at the voltage determined by the amplifier in its normal mode of operation. The difference between this output voltage and the output voltage when in normal operational mode, divided by the gain, represents the input offset voltage. With an older processing technology, the gain of the auto-zero stage may be on the order of 60 decibels (dB) or 1000 times. For 1 mV of offset and equal gain from the main and auxiliary inputs, there is an error of 1 μV at the input of the amplifier. If an attempt to reduce the charge injection error is made that reduces the effective gain to the auxiliary inputs, the offset increases proportionally. If the effective gain on the auxiliary inputs is reduced by a factor of a thousand in order to reduce the charge-injection caused offset to 1 μV, the open loop gain would have to be raised to 120 dB to keep the finite gain error at 1 μV. This is not practical for most applications.
Furthermore, there is broadband noise that is aliased down to baseband from the sampling operation. The noise has several facets, some of which are related. One is the noise of the amplifier in the total bandwidth of the auto-zero loop when it is brought down to baseband. The other noise is the
      kT    C  noise (also referred to as the kTC noise) of the hold capacitor at the auxiliary input of the amplifier, where k is the Boltzmann's Constant of approximately 1.3807×10−23 Joules per Kelvin (J K−1), T is the absolute temperature, and C is the capacitance of Chold. The kTC noise is quite large for values of hold capacitor that can be integrated on chip. For example, it is on the order of 20 μV for a 10 pF capacitor.
One approach to address the above issues is to use large capacitors. In some implementations, the capacitors may be external to the chip containing the amplification circuit. In others implementations the large area of the capacitors may be tolerated on the chip. The use of newer deep sub-micron processes makes this possible, although this approach is expensive because of the large chip area and the advanced process technology employed.
As discussed above, the prior art solutions require large on-chip area, external components, or expensive newer processing technology. All such solutions lead to higher product costs. Therefore, there is a need for a low-cost solution for processing very small baseband signals that addresses the issues of charge injection error, finite gain limitations and noise problems.